Integrated driver and related method

ABSTRACT

A driver circuit may include a first node, and a first circuit to generate on the first node an inverted replica of an input signal during driver switching between a first supply voltage and a first reference voltage, the inverted replica having a threshold voltage value based upon a second reference voltage greater than the first supply voltage. The driver circuit may include a cascode stage to be controlled by the second reference voltage and to be coupled between a second supply voltage and the first node, a delay circuit to generate a delayed replica of the input signal, an amplifier, and a switching network to couple the control terminal of the active load transistor to one of the first reference voltage and the first node based upon the input signal.

FIELD OF THE INVENTION

The present disclosure relates to drivers for capacitive loads and, more particularly, to single-ended or differential integrated architectures of drivers including voltage transistors.

BACKGROUND OF THE INVENTION

In the emerging technology known as “Silicon Photonics,” optical devices are integrated with electronic components. A classical “Silicon Photonics” application includes an optical transmitter and an optical receiver and is shown in FIG. 1. A high frequency data stream is transmitted by modulating a laser beam. By way of this beam modulation, an electrical data stream is converted into an optical data stream, which is more suitable for a long-distance low-loss data transmission.

Together with optical modulators, optical switches are commonly integrated with electrical functional circuits in a “Silicon Photonic” integrated circuit. Driving these optical modulators commonly may require both high voltage and high speed capabilities, with the constraint of driving large capacitive loads with very short rise and fall time constants.

An example of these functional circuits is the driver for the Mach Zehnder optoelectronic modulator of FIG. 2. An integrated Mach Zehnder (MZ) optoelectronic modulator is a combination of two directional couplers interconnected by two symmetrical silicon waveguides of a given length. The group velocity of the light into the two waveguides is controlled by the voltage polarization of two varicap diodes. The working principle of the MZ optoelectronic modulator is based on the constructive or destructive interference of the two separated paths as a consequence of the two different light group velocities.

In an MZ optoelectronic modulator, such a selective interference is used to modulate a continuous wave laser beam, but the same principle of constructive or destructive interference is commonly used also to perform other optical functions, like optical switches variable attenuators etc.

The driver is commonly required to have high current capability at high power efficiency. If C_(L) is the value of the capacitance of the MZ optoelectronic modulator, the required capability of the output driver may be approximated as follows:

${{Iout} = \frac{{C_{L} \cdot V}\; {out\_ peak}}{{tr}\left( {{or}\mspace{14mu} {tf}} \right)}},$

wherein tr and tf are the rise time and the fall time constants, respectively.

FIG. 3 shows a typical application. Due to high speed requirements, high-speed complementary metal-oxide-semiconductor (CMOS) technologies are commonly selected for the electronic part. These may include the draw-back of the voltage capability of the CMOS not being adequate for the voltage levels required for driving the MZ. For this reason, the driver is commonly required to drive the MZ optoelectronic modulator with a voltage swing exceeding the low voltage CMOS supply (V_(dd1)).

Prior art approaches may include high voltage level shifters to convert the CMOS digital input signals, that are constrained within a low voltage supply (Vdd1), to the output buffer voltage levels (Vout) required to drive the Mach Zehnder optoelectronic modulators, as schematically shown in FIG. 3. Moreover, prior art approaches may also have a fixed output swing without any possibility of adjusting the peak value of the output voltage Vout. U.S. Patent Application Publication No. 2009/0148094 to Kucharski et al. and U.S. Pat. No. 7,450,787 to Kucharski et al. disclose an optoelectronic device having a plurality of optical modulators and a plurality of distributed amplifiers, each electrically coupled to a respective optical modulator. U.S. Pat. No. 7,519,301 to Keil et al. discloses several circuit architectures of emitter follower-based or source follower-based drivers coupled to drive a Mach-Zehnder interferometer optical modulator. U.S. Pat. No. 7,199,617 discloses a level shifting device that can translate an input signal operating in a first voltage range to an output signal operating in other voltage ranges while using transistors rated to withstand the supply voltage.

SUMMARY OF THE INVENTION

The MZ optoelectronic modulator is indicated as an exemplary field of application of the applicant's disclosure, though the disclosed architectures may be used also for driving in general any kind of load, in particular other capacitive loads and not exclusively MZ optoelectronic modulators.

A flexible and cost efficient driver architecture may be adapted to drive capacitive loads, such as, for example, a Mach Zehnder optoelectronic modulator.

The driver architecture may comprise low-voltage transistors and may generate a high output driving voltage with a large swing without requiring a high-voltage level shifter. Single-ended and differential architectures of an integrated CMOS driver architecture, particularly suited to drive a substantially capacitive load, such as, for example, a Mach Zehnder optoelectronic modulator, are disclosed.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a basic high-level block diagram of an integrated optical transmitter and receiver, according to the prior art.

FIG. 2 illustrates two single-ended drivers for a Mach Zehnder optoelectronic modulator, according to the prior art.

FIG. 3 illustrates two single-ended drivers for a Mach Zehnder optoelectronic modulator with a fixed swing voltage, each having an input level shifter for increasing the voltage level of the differential input signal V_(IN)−V′_(IN), according to the prior art.

FIG. 4 is an embodiment of a single-ended CMOS driver architecture, according to the present invention.

FIG. 5 is an embodiment of a differential CMOS driver architecture, according to the present invention.

FIGS. 6 and 7 are graphs of the main signals of the CMOS driver architecture of FIG. 4.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

Fully integrated single-ended and differential CMOS driver architectures suitable for realizing high speed driving stages of different types of loads, in particular, of a Mach Zehnder optoelectronic modulator or of distributed fully differential Mach Zehnder optoelectronic modulators, which show a dominant capacitive load impedance, are shown.

FIGS. 4 and 5 show several embodiments of the single-ended and differential driver architectures, respectively. These architectures may be realized with any modern VLSI CMOS technology with the availability of dual gate oxide, and also with any modern BICMOS technology. The functioning of the driver architecture will be described while referring to the single-ended embodiment of FIG. 4, though the same considerations hold the same for the differential embodiment of FIG. 5.

An input inverter including the transistors M1-M2 (M′1-M′2) is driven by a CMOS compatible switching input signal VIN (V′IN). An integrated capacitor C (C′) is pre-charged at a desired voltage equal to Vref−Vth, wherein Vth is the threshold voltage of the cascode stage M3 (M′3), during one semi-period of the input square waves. In this way, the working point of the cascode stage M3 (M′3) switches from a deep cutoff condition to a functioning condition close to the interdiction region, in order to restore the initial charge of the capacitor C. Whenever the output of the inverter is switched to the low voltage supply Vdd1, the switching network M4, M5 and M6 (M′4, M′5 and M′6) is configured to force at a voltage close to Vdd1+Vref−Vth the gate of the single high-voltage transistor M9 (M′9), that functions as an active load, thus causing a great overdrive to the transistor M9 (M′9). The class AB amplifier, comprising by the three transistors M9, M8 and M7 (M′9, M′8 and M′7), ensures an output peak voltage equal to:

V_(dd1)+V_(ref)−V_(th)−V_(gs)M₉;

wherein V_(gs)M₉ is the gate-source voltage on the transistor M₉.

The V_(out) peak could also be programmable by adjusting the reference voltage V_(ref). This last feature may be desirable also for calibration purposes. The high efficiency (class AB) and cross-conduction of the three output transistors are optimized by calibrating the delay D (D′) according to the characteristics of the load in order to turn-on the transistors M₇ and M₈ (M′₇ and M′₈) only when M₉ (M′₉) has turned off. The two transistors M₅ (M′₅) and M₆ (M′₆) of the switching network are used to ensure a fast discharge of the gate-source capacitance of M₉ (M′₉).

The time graphs of FIGS. 6 and 7 show exemplary waveforms when the input voltage V_(in) (V′_(in)) switches between Vdd1 and ground. The shown exemplary pattern is made of positive and negative cycles whose duration is nT, wherein T is a known time unit interval, n assuming any integer value between 1 and N. The value of capacitors C (C′) is preferably chosen proportionally to N and according to the output ripple that may be tolerated on the output voltage V_(out) (V′_(out)).

When the input voltage V_(IN) (V′_(IN)) switches low, the voltage V_(A) (V′_(A)) switches high and the control terminal of the active load M₉ (M′₉) is disconnected from ground. Therefore, the switch M₄ (M′₄) turns on and the voltage V_(B) (V′_(B)) equals the voltage V_(A) (V′_(A)). After a certain time delay, determined by the delay D (D′), has elapsed, the transistor M₇ (M′₇) and thus also the transistor M₈ (M′₈) turn off. Accordingly, the output voltage V_(out) (V′_(out)) on the supplied capacitive load increases and becomes practically equal to the voltage V_(B) (V′_(B)), i.e. in practice, equal to the voltage V_(A) (V′_(A)) (i.e. determined by the reference voltage V_(ref) (V′_(ref))). During this phase, the supplied capacitive load is charged through the transistor M₉ (M′₉), thus the rise time t_(r) of the output voltage may be finely determined by fixing the resistance of the transistor M₉ (M′₉) in a conduction state.

When the input voltage V_(IN) (_(V′IN)) switches high, the voltage V_(A) (V′_(A)) switches low at a voltage slightly smaller than the reference voltage V_(ref) (V′_(ref)) in a time interval determined by the recovery time of the transistor M₁ (M′₁). Therefore, the transistors M₅ (M′₅) and M₆ (M′₆) of the switching network turn on, the control terminal of the active load M₉ (M′₉) is grounded and the transistor M₄ (M′₄) is off. After a certain time delay, determined by the delay D (D′), has elapsed, the transistor M₇ (M′₇) and thus also the transistor M₈ (M′₈) turn on, thus the output voltage V_(out) on the supplied capacitive load decreases. During this phase, the supplied capacitive load is discharged through the transistors M₈ (M′₈) and M₇ (M′₇), thus the fall time t_(f) of the output voltage may be finely determined by fixing the resistance in conduction state of these transistors.

The transistors may be sized to get the desired output peak current and the desired rise and fall time simply by adjusting the aspect ratio of the transistors M₇(M′₇) and M₉ (M′₉) and the delay D (D′). In order to limit the total in-out current capability, a series resistor R may be connected in series to the supplied load, as shown in FIG. 4.

The disclosed approach does not need any type of level shifter to adapt the low voltage input signal to the output buffer, furthermore it allows Vout peak programmability through a reference voltage V_(ref), that may be easily controlled by a digital-to-analog converter. Moreover, the disclosed architectures have a low power consumption and a high power efficiency. The claims as filed are integral part of this description and are herein incorporated by reference. 

1-9. (canceled)
 10. A method of generating an output signal switching between a second supply voltage and a first reference voltage, starting from an input signal switching between a first supply voltage and the first reference voltage using an amplifier, the amplifier having a first switch to be coupled to the first reference voltage and to be controlled by a delayed replica of the input signal, a second switch to be controlled by the first supply voltage and comprising a conduction terminal to define an output terminal, and an active load switch to be coupled between the output terminal and the second supply voltage and comprising a control terminal, the method comprising: generating an inverted replica of the input signal having a threshold voltage based upon a second reference voltage greater than the first supply voltage; controlling the first switch with the delayed replica of the input signal; controlling the second switch with the first supply voltage; and coupling the control node of the active load switch to at least one of the first reference voltage and the inverted replica of the input signal based upon on a level of the input signal.
 11. The method of claim 10 wherein the input signal comprises a square-wave input signal; and wherein the output signal comprises a square-wave output signal.
 12. The method of claim 10 wherein the conduction terminal of the second switch is not in common with the first switch.
 13. The method of claim 10 wherein the active load switch comprises a high-voltage active load switch.
 14. A method of operating a driver circuit comprising: generating on a first node an inverted replica of an input signal during driver switching between a first supply voltage and a first reference voltage, the inverted replica having a threshold voltage value based upon a second reference voltage greater than the first supply voltage; operating a cascode stage controlled by the second reference voltage and coupled between a second supply voltage and said first node; generating a delayed replica of the input signal; operating an amplifier comprising a first switch coupled to the first reference voltage and controlled by the delayed replica of the input signal, a second switch controlled by the first supply voltage and comprising a conduction terminal to define an output terminal, and an active load switch coupled between the output terminal and the second supply voltage and comprising a control terminal; and coupling said control terminal of said active load switch to at least one of the first reference voltage and said first node based upon the input signal.
 15. The method of claim 14 wherein the input signal comprises a square-wave input signal; and wherein coupling said control terminal of said active load switch is based upon a level of the square-wave input signal.
 16. The method of claim 14 wherein generating on the first node the inverted replica comprises: inverting the input signal using a plurality of switches having a common node; and operating a boost capacitor coupled between said common node of said plurality of switches and said first node to be charged at a voltage to keep said cascode stage in a functioning mode.
 17. The method of claim 14 wherein generating the delayed replica comprises generates the delayed replica to turn on said first switch after said active load switch has turned off.
 18. The method of claim 14 wherein said first switch, and said second switch, each comprises a low-voltage metal-oxide-semiconductor field-effect transistor (MOSFET).
 19. The method of claim 14 wherein said conduction terminal of said second switch is not in common with said first switch.
 20. The method of claim 14 wherein said active load switch comprises a high-voltage active load switch.
 21. The method of claim 14 wherein said amplifier comprises a class AB amplifier.
 22. A method of controlling a Mach-Zender optical modulator comprising: generating on a first node an inverted replica of an input signal during driver switching between a first supply voltage and a first reference voltage, the inverted replica having a threshold voltage value based upon a second reference voltage greater than the first supply voltage; operating a cascode stage controlled by the second reference voltage and coupled between a second supply voltage and said first node; generating a delayed replica of the input signal; operating an amplifier comprising a first switch coupled to the first reference voltage and controlled by the delayed replica of the input signal, a second switch controlled by the first supply voltage and comprising a conduction terminal to define an output terminal coupled to the Mach-Zender optical modulator, and an active load switch coupled between the output terminal and the second supply voltage and comprising a control terminal; and coupling said control terminal of said active load switch to at least one of the first reference voltage and said first node based upon the input signal.
 23. The method of claim 22 wherein the input signal comprises a square-wave input signal; and wherein coupling said control terminal of said active load switch is based upon a level of the square-wave input signal.
 24. The method of claim 22 wherein generating on the first node the inverted replica comprises: inverting the input signal using a plurality of switches having a common node; and operating a boost capacitor coupled between said common node of said plurality of switches and said first node to be charged at a voltage to keep said cascode stage in a functioning mode.
 25. The method of claim 22 wherein generating the delayed replica comprises generates the delayed replica to turn on said first switch after said active load switch has turned off.
 26. The method of claim 22 wherein said first switch, and said second switch, each comprises a low-voltage metal-oxide-semiconductor field-effect transistor (MOSFET).
 27. The method of claim 22 wherein said conduction terminal of said second switch is not in common with said first switch.
 28. The method of claim 22 wherein said active load switch comprises a high-voltage active load switch.
 29. The method of claim 22 wherein said amplifier comprises a class AB amplifier. 